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AURORA BOREALIS seen from Churchill, Manitoba, Canada. Ionospheric scintillation research can benefit from this new method. (Photo: Aiden Morrison) Photo: Canadian Armed Forces By Aiden Morrison, University of Calgary Two broad user groups will find important consequences in this article: Time synchronization and test equipment manufacturers, whose GPS-disciplined oscillators have excellent long-term performance but short- to medium-term behavior limited by the quality, and therefore cost, of the integrated quartz device. This article portends a family of devices delivering oven-controlled crystal oscillator (OCXO) performance down to the 10-millisecond level, with an oscillator costing pennies, rather than tens or hundreds of dollars. Applications include ionospheric scintillation research (above). High-performance receiver manufacturers who design products for high-dynamic or high-vibration environments (see cover) where the contribution of phase noise from the local oscillator to velocity error cannot be ignored. In these areas, the strategy outlined here would produce equipment that can perform to higher specifications with the same or a lower-cost oscillator. The trade-off requires two tracking channels per satellite signal, but this should not pose a problem. At ION GNSS 2009, manufacturers showed receivers with 226 tracking channels. There are currently only 75 live signals in the sky, including all of GPSL1/L2/L5 and GLONASS L1/L2. — Gérard Lachapelle If the channel data within a GNSS receiver is handled in an effective manner, it is possible to form meaningful estimates of the local-oscillator phase deviations on timescales of 10 milliseconds (ms) or less. Moreover, if certain criteria are met, these estimates will be available with related uncertainties similar to the deviations produced by a typical oven-controlled crystal oscillator (OCXO). The processing delay required to form this estimate is limited to between 10 and 20 ms. In short, it becomes possible in near-real-time to remove the majority of the phase noise of a local oscillator that possesses short-term instability worse than an OCXO, using standalone GNSS. This represents both a new method to accurately determine the Allan deviation of a local oscillator at time scales previously impractical to assess using a conventional GNSS receiver, and the potential for the reduction in observable Doppler uncertainty at the output of the receiver, as well as ionospheric scintillation detection not reliant on an expensive local OCXO. Concept. Inside a typical GNSS receiver, the estimate of the error in the local oscillator is formed as a component of the navigation solution, which is in turn based on the output of each satellite-tracking channel propagating its estimate of carrier and code measurements to a common future point. While this method of ensuring simultaneous measurements is necessary, it regrettably limits the resolution with which the noise of the local oscillator can be quantified, due to the scaling of non-simultaneous samples of local oscillator noise through the measurement propagation process. To bypass these shortcomings requires a method of coherently gathering information about the phase change in the local oscillator across all available satellite signals: to use the same samples simultaneously for all satellites in view to estimate the center-point phase error common across the visible constellation. To explain how this is feasible, we must first understand the limitations imposed by the conventional receiver architecture, with respect to accurately estimating short-term oscillator behavior, and subsequently to determine the potential pitfalls of the proposed modifications, including processing delays needed for bit wipe-off, expected observation noise, and user dynamics effects. Typical Receiver Shortfalls In a typical receiver, while information about local time offset and local oscillator frequency bias may be recovered, information about phase noise in the local oscillator is distorted and discarded, as a consequence of scaling non-simultaneous observations to a common epoch. As shown in FIGURE 1, coherent summation intervals in a receiver are used to approximate values of the phase error, including oscillator phase, measured at the non-simultaneous interval centrers in each channel, which are then propagated to a common navigation solution epoch. Each channel will intrinsically contain a partially overlapping midpoint estimate of oscillator noise over the coherent summation interval that will then be scaled by the process of extrapolation. As these estimates are scaled and partially overlapping, they do not make optimal use of the information known about the effects of the local oscillator, and form a poor basis for estimating the contributions of this device to the uncertainty in the channel measurements. As shown in Figure 1, the phase error measured in each channel will be distorted by an over unity scaling factor. FIGURE 1. Propagation and scaling of phase estimates withina typical receiver. Depending on implementation decisions made by the designers of a given GNSS system, the average value of the propagation interval relative to the bit period will have different expected values. Assuming the destination epoch is the immediate end of the furthest advanced (most delayed) satellite bitstream, and that integration is carried out over full bit periods, the minimum propagation interval for this satellite would be ½-bit period. For the average satellite however, the propagation delay would be this ½-bit period plus the mean skew between the furthest satellite and the bitstreams of other space vehicles. Ignoring further skew effects due to the clock errors within the satellites, which are typically limited well below the ms level, the skew between highest and lowest elevation GPS satellites for a user on the surface of earth would be approximately 10 ms. The average value of this skew due to ranging change over orbit, assuming an even distribution of satellites in the sky at different elevation angles, would therefore be 5 ms. Combining the minimum value of the skew interval with the minimum propagation interval of the most delayed satellite yields a total average propagation interval of 15 ms. In turn, this gives a typical scaling factor of 1.75, used from this point forward when referring to the effects of scaling this quantity. Proposed Implementation Overcoming limitations of a typical receiver requires recording the approximate bit-timing and history of each tracked satellite as well as a short segment of past samples. This retained data guarantees that the bit-period boundaries of the satellites will not pose an obstacle to forming common N-ms coherent periods between all visible satellites, over which simultaneous integration may proceed by wiping off bit transitions. Using this approach as shown in FIGURE 2, all available constellation signal power is used to estimate a single parameter, namely the epoch-to-epoch phase change in the local oscillator. FIGURE 2. Common intervals over which to accurately estimate local oscillator phase changes. Having viewed the existence of these common periods, it becomes evident that it is conceptually possible to form time-synchronized estimates of the phase contribution of the common system oscillator alternately across one N-ms time slice, then the next, in turn forming an unb roken time series of estimates of the phase change of the system oscillator. Forming the difference between the adjacent discriminator outputs will provide the following information: The ΔEps (change in the noise term in the local loop) The ΔOsc (change in the phase of the local oscillator, the parameter of interest) The ΔDyn (change in the untracked/residual of real and apparent dynamics of the local loop/estimator) Noticing that term 1 may be considered entirely independent across independent PRNs (GPS, Galileo, Compass) or frequency channels (GLONASS), and that the value of term 3 over a 10-ms period is expected to be small over these short intervals, it becomes obvious that term 2 can be recovered from the available information. To determine the weighting for each satellite channel, the variance of the output of the discriminator is needed. Performance Determination To allow the realistic weighting of discriminator output deltas, it becomes desirable to estimate at very short time intervals the variance of the output of the phase discriminator. In the case of a 2-quadrant arctangent discriminator, this means one wishes to quantify the variance Letting Q/I 5 Z, recall that if Y 5 aX then Applying this to the variance of the input to the arctangent discriminator in terms of the in phase and quadrature accumulators, this would give Rather than proceed with a direct evaluation from this point onward to determine the expression for the variance at the output of the discriminator, it is convenient to recognize that simpler alternatives exist since The implication is that since the slope of the arctangent transfer function is very nearly equal to 1 in the central, typical operating region, and universally less than 1 outside of this region, it is easy to recognize that the variance at the output of the arctangent discriminator is universally less than that at the input, and can be pessimistically quantified as the variance of the input, or  σ2(Z). This assumption has been verified by simulation, its result shown in FIGURE 3, where the response has been shown after taking into account the effect of operating at a point anywhere in the range ±45 degrees. While the consequence of the simplification of the variance expression is an exaggeration of discriminator output variance, FIGURE 4 shows output variance is well bounded by the estimate, and within a small margin of error for strong signals. FIGURE 3. Predicted variances at the output of the ATAN2discriminator versus C/N0. FIGURE 4. Difference between actual and predicted variance at output of discriminator. The gap between real and predicted output variance may also be narrowed in cases where Q>I by using a type of discriminator which interchanges Q and I in this case and adds an appropriate angular offset to the output as Proceeding in this vein, the next required parameter is the normalized variance of the in-phase and quadrature arms. The carrier amplitude A can be roughly approximated as Resulting in a carrier power C Further, the noise power is given as Expressing bandwidth B as the inverse of the coherent integration time, and rearranging now gives noise density N0 as Combining this expression, and the one previously given for the carrier power C results in the following expression for the carrier to noise density ratio: This latest expression can be rearranged to find the desired variance term. Assuming the 10-ms coherent integration time discussed earlier is used, this yields Normalizing for the carrier amplitude gives the normalized variance in terms of radians squared: In any situation where the carrier is sufficiently strong to be tracked, it is likely that the carrier power term employed above can be gathered from the immediate I and Q values, ignoring the contribution of the noise term to its magnitude. Oscillator Phase Effect. Determining the expected magnitude of the local oscillator phase deviation requires only three steps, assuming that certain criteria can be met. The first requirement is that the averaging times in question must be short relative to the duration, at which processes other than white phase and flicker phase modulation begin to dominate the noise characteristics of the oscillator. Typically the crossover point between the dominance of these processes and others is above 1 s in averaging interval length, when quartz oscillators are concerned. Since this article discusses a specific implementation interval of 10 ms within systems expected to be using quartz oscillators, it is reasonable to assume that this constraint will be met. The second requirement is that the Allan deviation of the given system oscillator must be known for at least one averaging interval within the region of interest. Since the Allan deviation follows a linear slope of -1 with respect to averaging interval on a log-log scale within the white-phase noise region, this single value will allow an accurate prediction of the Allan deviation at any other point on the interval and, in turn, of the phase uncertainty at the 10 ms averaging interval level. Letting σΔ(τ) represent the Allan deviation at a specific averaging interval, recall that this quantity is the midpoint average of the standard deviation of fractional frequency error over the averaging interval τ. Scaling this quantity by a frequency of interest results in the standard deviation of the absolute frequency error on the averaging interval: By integrating this average difference in frequency deviations over the coherent period of interest, one obtains a measure of the standard deviation in degrees, of a signal generated by this reference: Note that the averaging interval τ must be identical to the coherent integration time. Turning to a practical example, if the oscillator in question has a 1 s Allan Deviation of 1 part per hundred billion (1 in 1011), a stability value between that of an OCXO and microcomputer compensated crystal oscillator (MCXO) standard, and shown to be somewhat pessimistic, this would scale linearly to be 1e-9 at a 10-ms averaging interval, under the previous assumption that the oscillator uncertainty is dominated by the white phase-noise term at these intervals. Also, for illustration purposes, if one assumes the carrier of interest to be the nominal GPS L1 carrier, the uncertainty in the local carrier replica due to the local oscillator over a 10-ms coherent integration time becomes When stated in a more readily digested format, this represents roughly 15 centimeter/second in the line-of-sight velocity uncertainty. In an operating receiver, two additional factors modify this effect. The first is the previously discussed scaling effect that will tend to exaggerate this effect by a typical factor of 1.75, as previously discussed. The second factor is that this noise contribution is filtered by the bandwidth-limiting effects of the local loop filter, producing a modification to the noise affecting velocity estimates, as well as reduced information about the behaviour of the local oscillator. Impact of Apparent Dynamics. When considering the error sources within the system, it is important to realize which individual sources of error will contribute to estimation errors, and which will not. One area of potential concern would appear to be the errors in the satellite ephemerides, encompassing both the satellite-orbit trajectory misrepresentation and the satellite clock error. While the errors in the satellite ephemerides are of concern for point positioning, they are not of consequence to this application, as the apparent error introduced by a deviation of the true orbit from that expressed in the broadcast orbital parameters does not affect the tracking of that satellite at the loop level. Additionally, while the satellite clock will add uncertainty to the epoch-to-epoch phase change within each channel independently, the magnitude of this change is minimal relative to the contribution of uncertainty due to the variance at the output of the discriminator guaranteed by the low carrier-to-noise density ratio of a received GNSS signal. Since this contribution is uncorrelated between satellites and relatively small compared to other noise contributions affecting these measurements, even when compared to the soon-to-be-discontinued Uragan GLONASS satellites that had generally less stable onboard clocks, it is likely safe to ignore. When compared to the more stable oscillators aboard GPS or GLONASS-M satellites, it is a reasonable assumption that this will be a dismissible contribution to received signal-phase uncertainty change. While atmospheric effects present an obstacle which will directly affect the epoch-to-epoch output of the discriminators, it is believed that under conditions that do not include the effects of ionospheric scintillation the majority of the contribution of apparent dynamics due to atmospheric changes will have a power spectral density (PSD) heavily concentrated below a fraction of 1 Hz. The consequence of this concentration is that the tracking loops will remove the vast majority of this contribution, and that the difference operator that will be applied between adjacent phase measurements, as in the case of dynamics, will nullify the majority of the remaining influence. Impact of Real Dynamics. Real dynamics present constraints on performance, as do any tracking loop transients. For example, a low-bandwidth loop-tracking dynamics will have long-lasting transients of a magnitude significant relative to levels of local oscillator noise. For this reason it is necessary to adopt a strategy of using the epoch-to-epoch change in the discriminator as the figure of interest, as opposed to the absolute error-value output at each epoch. This can reasonably be expected to remove the vast majority of the effects of dynamics of the user on the solution. To validate this assumption under typical conditions calls for a short verification example. Assuming the use of a second-order phase-locked loop (PLL) for carrier tracking, with a 10-Hz loop bandwidth the effects of dynamics on the loop are given by these equations: Letting Bn be 10 Hz, one can write Recall that the dynamic tracking error in a second-order tracking loop is given by Given the choices above, this would result in a constant offset of 0.00281 cycles, or 1.011 degrees of constant tracking error due to dynamics, following from the relation between line-of-sight acceleration and loop bandwidth to tracking error. Since this constant bias will be eliminated by the difference operator discussed earlier, it is necessary to examine higher order dynamics. Further, if one used a coherent integration interval of 10 ms as assumed earlier, and let the dynamics of interest be a jerk of 1 g/s, this results in a midpoint average of 0.005 g on this interval: Substituting this result into equation 16 produces the associated change in dynamic error over the integration interval, which is in this case: This value will be kept in mind when evaluating capabilities of the estimation approach to determine when it will be of consequence. As the estimation process will be run after a short delay, an existing estimate of platform dynamics could form the basis of a smoothing strategy to reduce this dynamic contribution further. Estimated Capabilities In the absence of the influence of any unmodeled effects, the expected performance of this method is dependent on only the number of satellite observables and their respective C/N0 ratios. Across each of these scenarios we assume for simplicity’s sake that each satellite in view is received at a common C/N0 ratio and over a common integration period of 10 ms. If the assumption of minimal dynamic influences is met, the situation at hand becomes one in which multiple measures of a single quantity are present, each containing independent (due to CDMA or FDMA channel separation) noise influences with a nearly zero mean. When one can express the available data form: x[n] = R + w[n] where x[n] is the nth channel discriminator delta which includes the desired measure of the local oscillator delta (R), as well as w[n], a strong, nearly white-noise component, there are multiple approaches for the estimation of R. The straightforward solution to estimate R in this case is to use the predicted variances of each measure to serve as an inverse weighting to the contribution of each individual term, followed by normalization by the total variance, as expressed by Now, since it is desired to bound the uncertainty of the estimate of R, the variance of this quantity should also be noted. This uncertainty can be determined as To determine the performance of the estimation method for a given constellation configuration, with specific power levels and available carrier signals, it is necessary to utilize the predicted variances plotted in Figure 3 as inputs to equations 20 and 21. To provide numerical examples of the performance of this method, three scenarios span the expected range of performance. Scenario 1 is intended to be char-acteristic of that visible to a single-freq-uency GPS user under slight attenuation. It is assumed that 12 single-frequency satellites are visible at a common C/N0 of 36 dB-Hz, yielding from the simulation curves a value for each channel of 0.0265 rad2. When substituted into equation 24, this predicts an estimation uncertainty of This is a level of estimation uncertainty similar to that assumed to be intrinsic to the local oscillator in the previous section. The result implies that with this minimally powerful set of satellites, it becomes possible to quantify the behavior of the local oscillator with a level of uncertainty commensurate with the actual uncertainty in the oscillator over the 10 ms averaging interval. Consequentially, this indicates that the Allan deviation of this system oscillator could be wholly evaluated under these conditions at any interval of 10 ms or longer. Further, if the system oscillator were in fact the less stable MCXO from the resource above, this estimate uncertainty would be significantly lower than the actual uncertainty intrinsic to the oscillator, providing an opportunity to “clean” the velocity measurements. Scenario 2 is intended to be characteristic of a near future multi-constellation single-frequency receiver. It is assumed that eight satellites from three constellations are visible on a single frequency each, with a common C/N0 of 42 dB-Hz, yielding a value for each channel of 6.4e-3 rad2, leading to an estimation uncertainty of Scenario 3 is intended to serve as an optimistic scenario involving a future multi-frequency, multi-constellation receiver. It is assumed that nine future satellites are available from each of three constellations, each with four independent carriers, all received at 45 dB-Hz, yielding a value for each channel of 3.2e-3 rad2, leading to an estimation uncertainty of Application to Observations The theoretical benefit of subtracting these phase changes from the measurements of an individual loop prior to propagating that measurement to the common position solution epoch ranges from moderate to very high depending on the satellite timing skew relative to the solution point. The most beneficial scenario is total elimination of oscillator noise effects (within the uncertainty of the estimate), which is experienced in the special case (Case A, FIGURE 5), where the bit period of a given satellite falls entirely over two of the 10-ms subsections. The uncertainty would increase to 2x the level of uncertainty in the estimate in the special case (Case B) where the satellite bit period straddles one full 10-ms period and two 5-ms halves of adjacent periods, and would lie somewhere between 1 and 2 times the level of uncertainty for the general case where three subintervals are covered, yet the bit period is not centered (Case C). FIGURE 5. Special cases of oscillator estimate versus bit-period alignment. While the application to observations of the predicted oscillator phase changes between integration intervals does not appear immediately useful for high-end receiver users with the exception of those in high-vibration or scintillation-detection applications, it could be applied to consumer-grade receivers to facilitate the use of inexpensive system clocks while providing observables with error levels as low as those provided by much more expensive receivers incorporating ovenized frequency references. Further Points While the chosen coherent integration period may be lengthened to increase the certainty of the measurement from a noise averaging perspective, this modification risks degrading the usefulness of said measurement due to dynamics sensitivities. Additionally, as the coherent integration time is increased, the granularity with which the pre-propagation oscillator contribution may be removed from an individual loop will be reduced. While this may be useful in cases of very low dynamics where the system is intended to estimate phase errors in a local oscillator with high certainty, it would be of little use if one desires to provide low-noise observables at the output. For this reason, it is recommended that increases in coherent integration time be approached with caution, and extra thought be given to use of dynamics estimation techniques such as smoothing, via use of the subsequent n-ms segment in the formation of the estimate of dynamics for the “current” segment. This carries the penalty of increased processing latency, but could greatly reduce dynamics effects by enabling their more reliable excision from the desired phase-delta measurements. Acknowledgments The author thanks his supervisors, Gerard Lachapelle and Elizabeth Cannon, and the Natural Sciences and Engineering Research Council of Canada, the Alberta Informatics Circle of Research Excellence, the Canadian Northern Studies Trust, the Association of Canadian Universities for Northern Studies, and Environment Canada for financial and logistical support. AIDEN MORRISON is a Ph.D. candidate in the Position, Location, and Navigation (PLAN) Group, Department of Geomatics Engineering, Schulich School of Engineering at the University of Calgary, where he has developed a software-defined GPS/GLONASS receiver for his research.

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Digipower tc-500 travel charger 4.2/8 4vdc 0.75a used battery po.90w-lt02 ac adapter 19vdc 4.74a replacement power supply laptop,this circuit shows the overload protection of the transformer which simply cuts the load through a relay if an overload condition occurs,is someone stealing your bandwidth.macintosh m3037 ac adapter 24vdc 1.87a 45w powerbook mac laptop,insignia e-awb135-090a ac adapter 9v 1.5a switching power supply,dve dsc-6pfa-05 fus 070070 ac adapter 7v 0.7a switching power su,wahl db06-3.2-100 ac adapter 3.2vdc 100ma class 2 transformer.illum fx fsy050250uu0l-6 ac adapter 5vdc 2.5a used -(+) 1x3.5x9m,phihong psc30u-120 ac adapter 12vdc 2.5a extern hdd lcd monitor,neuling mw1p045fv reverse voltage ac converter foriegn 45w 230v,opti pa-225 ac adapter +5vdc +12vdc 4pins switching power supply,gps l1 gps l2 gps l3 gps l4 gps l5 glonass l1 glonass l2 lojack.cincon tr36a-13 ac adapter 13.5v dc 2.4a power supply,90 %)software update via internet for new types (optionally available)this jammer is designed for the use in situations where it is necessary to inspect a parked car,f10603-c ac adapter 12v dc 5a used 2.5 x 5.3 x 12.1 mm,auto no break power supply control,koolatron abc-1 ac adapter 13v dc 65w used battery charger 120v,it can be configured by using given command.oem ads18b-w 220082 ac adapter 22vdc 818ma new -(+)- 3x6.5mm ite.mobile jammer india deals in portable mobile jammer, gps blocker ,siemens 69873 s1 ac adapter optiset rolm optiset e power supply,yardworks 24990 ac adapter 24vdc 1.8a battery charger used power,jvc aa-v15u ac power adapter 8.5v 1.3a 23w battery charger,chicony w10-040n1a ac adapter 19vdc 2.15a 40w used -(+) 1.5x5.5x,apple powerbook duo aa19200 ac adapter 24vdc 1.5a used 3.5 mm si,ast ad-5019 ac adapter 19v 2.63a used 90 degree right angle pin,darelectro da-1 ac adapter 9.6vdc 200ma used +(-) 2x5.5x10mm rou.eng 3a-161wp05 ac adapter 5vdc 2.6a -(+) 2.5x5.5mm 100vac switch,battery technology mc-ps/g3 ac adapter 24vdc 2.3a 5w used female.linearity lad6019ab5 ac adapter 12vdc 5a used 2.5 x 5.4 x 10.2 m.linearity lad1512d52 ac adapter 5vdc 2a used -(+) 1.1x3.5mm roun.premium power 298239-001 ac adapter 19v 3.42a used 2.5 x 5.4 x 1.apple macintosh m4402 24vdc 1.875a 3.5mm 45w ite power supply,bestec bpa-301-12 ac adapter 12vdc 2.5a used 3 pin 9mm mini din.liteon pa-1650-02 ac adapter 19v dc 3.42a used 2x5.5x9.7mm,stancor sta-4190d ac adapter 9vac 500ma used 2x5.4mm straight ro.black & decker s036c 5102293-10 ac adapter 5.5vac 130ma used 2.5.milwaukee 48-59-2401 12vdc lithium ion battery charger used.olympus ps-bcm2 bcm-2 li-on battery charger used 8.35vdc 400ma 1,sony adp-120mb ac adapter 19.5vdc 6.15a used -(+) 1x4.5x6.3mm,this cell phone jammer is not applicable for use in europe.finecom ah-v420u ac adapter 12v 3.5a power supply,sony ac-lm5a ac adapter 4.2vdc 1.7a used camera camcorder charge,simple mobile jammer circuit diagram,ihome kss24-075-2500u ac adapter 7.5vdc 2500ma used -(+) 2x5.5x1,9 v block battery or external adapter,component telephone u070050d ac adapter 7vdc 500ma used -(+) 1x3.artesyn ssl40-3360 ac adapter +48vdc 0.625a used 3pin din power.shenzhen sun-1200250b3 ac adapter 12vdc 2.5a used -(+) 2x5.5x12m.astrodyne spu16a-105 ac adapter 12vdc 1.25a -(+)- 2x5.5mm switch.realistic 20-189a ac adapter 5.8vdc 85ma used +(-) 2x5.5mm batte,hp ppp017l ac adapter 18.5vdc 6.5a 5x7.4mm 120w pa-1121-12hc 391,ksah2400200t1m2 ac adapter 24vdc 2a used -(+) 2.5x5.5mm round ba,black & decker 680986-28 ac adapter 6.5vac 125va used power supp.building material and construction methods,compaq presario ppp005l ac adapter 18.5vdc 2.7a for laptop.super mobilline 12326 mpc 24vdc 5a charger 3pin xlr male used de.

Cisco systems adp-33ab ac adapter +5v +12v -12v dc 4a 1a 100ma.intermediate frequency(if) section and the radio frequency transmitter module(rft),v-2833 2.8vdc 165ma class 2 battery charger used 120vac 60hz 5w.xenotronixmhtx-7 nimh battery charger class 2 nickel metal hyd.maisto dpx351326 ac adapter 12vdc 200ma used 2pin molex 120vac p,digipower tc-3000 1 hour universal battery charger,apple m4551 studio display 24v dc 1.875a 45w used power supply.designed for high selectivity and low false alarm are implemented,fellowes 1482-12-1700d ac adapter 12vdc 1.7a used 90° -(+) 2.5x5.netgear dsa-12w-05 fus ac adapter 330-10095-01 7.5v 1a power sup.motorola spn4509a ac dc adapter 5.9v 400ma cell phone power supp.thomson 5-2608 ac adapter 9vdc 500ma used -(+) 2x5.5x9mm round b,sears craftsman 974775-001 battery charger 12vdc 1.8a 9.6v used.advent 35-12-200c ac dc adapter 12v 100ma power supply,the first types are usually smaller devices that block the signals coming from cell phone towers to individual cell phones,macintosh m4328 ac adapter 24.5vdc 2.65a powerbook 2400c 65w pow,seiko sii pw-0006-u1 ac adapter 6vdc 1.5a +(-) 3x6.5mm 120vac cl,hp 0950-2852 class 2 battery charger nicd nimh usa canada,hp 463554-002 ac adapter 19v dc 4.74a power supply.ac adapter 9vdc 500ma - ---c--- + used 2.3 x 5.4 x 11 mm straigh,mw mw1085vg ac adapter 10vdc 850ma new +(-)2x5.5x9mm round ba,samsung aa-e7 ac dc adapter 8.4v 1.5a power supply for camcorder,the pki 6160 is the most powerful version of our range of cellular phone breakers,cardio control sm-t13-04 ac adapter 12vdc 100ma used -(+)-.motorola psm4250a ac adapter 4.4vdc 1.5a used cellphone charger,motorola psm5091a ac adapter 6.25vdc 350ma power supply,a wide variety of custom jammers options are available to you,li shin lse9802a1240 ac adapter 12vdc 3.33a 40w round barrel.d-link van90c-480b ac adapter 48vdc 1.45a -(+) 2x5.5mm 100-240va.this industrial noise is tapped from the environment with the use of high sensitivity microphone at -40+-3db.adp da-30e12 ac adapter 12vdc 2.5a new 2.2 x 5.5 x 10 mm straigh,mastercraft maximum dc18us21-60 28vdc 2a class 2 battery charger.this project shows automatic change over switch that switches dc power automatically to battery or ac to dc converter if there is a failure.powerup g54-41244 universal notebook ac adapter 90w 20v 24v 4.5a.upon activation of the mobile jammer,therefore the pki 6140 is an indispensable tool to protect government buildings.altec lansing mau48-15-800d1 ac adapter 15vdc 800ma -(+) 2x5.5mm,fsp group inc fsp180-aaan1 ac adapter 24vdc 7.5a loto power supp,conswise kss06-0601000d ac adapter 6v dc 1000ma used,delta electronics adp-10ub ac adapter 5v 2a used -(+)- 3.3x5.5mm,delta sadp-135eb b ac adapter 19vdc 7.1a used 2.5x5.5x11mm power.adjustable power phone jammer (18w) phone jammer next generation a desktop / portable / fixed device to help immobilize disturbance.where shall the system be used,compaq pa-1900-05c1 acadapter 18.5vdc 4.9a 1.7x4.8mm -(+)- bul.biogenik 3ds/dsi ac adapter used 4.6v 1a car charger for nintend,ar 35-12-150 ac dc adapter 12v 150ma transmitter's power supply..

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